1. Field of the Invention
This invention relates generally to digital data communications, and more specifically to methods and apparatus for performing GMSK (Gaussian minimum shift keying) modulation in a digital communications system.
2. Description of the Prior Art
Several types of binary modulation techniques are known in the digital communications art. Frequency shift keying (FSK), phase shift keying (PSK), quadrature phase shift keying (QPSK), offset quadrature phase shift keying (OQPSK), and minimum shift keying (MSK) techniques, to name a few, have been employed in various applications in order to maximize either transmitted power efficiency or communications channel bandwidth efficiency or, in some cases, both of these critical communications system performance indicators. It is to be appreciated that, in particular, MSK modulation may be described from either a frequency modulation or phase modulation point of view. Specifically, it is known that MSK modulation is basically a coherently orthogonal, continuous phase, FSK modulation technique whereby the frequency deviation between the two discrete signaling frequencies equals one half of the data bit rate. Alternatively, it is also known that MSK modulation may be viewed as an OQPSK modulation technique with sinusoidal pulse weighting, i.e., the input data bits are represented by sinusoidal pulses instead of rectangular pulses. Nonetheless, from whichever point of view MSK modulation is described, the resulting output transmitted signal exhibits a constant envelope and continuity of phase in the RF carrier at each data bit transition.
Another binary modulation technique known as Gaussian minimum shift keying (GMSK) is also known in the relevant art. A modulator employing the GMSK binary modulation technique basically consists of an MSK modulator with a Gaussian low pass filter operatively coupled to the input of the MSK modulator. FIG. 1 illustrates one such conventional GMSK modulator. The GMSK modulator illustrated in FIG. 1 is described, in part, in the text entitled Wireless Digital Communications by Dr. Kamilo Feher, pages 164-175, as well as, in part, in U.S. Pat. No. 4,339,724 to Feher and U.S. Pat. No. 4,567,602 to Kato et. al. It is to be appreciated that the GMSK modulator illustrated in FIG. 1 represents the MSK modulation portion of the system from the above-mentioned quadrature phase modulation point of view.
Specifically, the conventional GMSK modulator illustrated in FIG. 1 includes a Gaussian low pass filter 2 operatively coupled to a non-return-to-zero (NRZ) bit stream source, an integrator 4 operatively coupled to a Gaussian low pass filter 2, and a signal splitter 6 operatively coupled to the integrator 4. The conventional GMSK modulator also includes a cosine look-up table 8 operatively coupled to the signal splitter 6, a sine look-up table 10 also operatively coupled to the signal splitter 6, a local oscillator 12, an in-phase mixer 14 operatively coupled to the local oscillator 12 and the cosine look-up table 8, a 90 degree phase shifter 16 operatively coupled to the local oscillator 12, a quadrature mixer 18 operatively coupled to the 90 degree phase shifter 16 and the sine look-up table 10, and a summer operatively coupled to the in-phase mixer 14 and the quadrature mixer 18.
The conventional GMSK modulator illustrated in FIG. 1 basically operates in the following manner. A standard NRZ input data bit stream, including individual bits having a value of either +1 or -1, corresponding to the information to be transmitted, is presented to the Gaussian low pass filter 2. An NRZ input bit stream may be generally represented as: ##EQU1## where a.sub.n =+/-1, T.sub.B is the bit interval, and .pi.(t/T.sub.B) is a rectangular pulse, where .pi.(t/T.sub.B) is represented as: ##EQU2## It is to be appreciated that the Gaussian low pass filter 2 is included as a pre-modulator filtering stage and serves the purpose of reducing the bandwidth of the main lobe and spectral density of the sidelobes associated with the transmitted GMSK output signal (i.e., bandwidth limiting the transmitted output signal). The frequency response of a Gaussian low pass filter is generally represented as: ##EQU3## where B represents the 3dB bandwidth of the Gaussian low pass filter.
The output signal from the Gaussian low pass filter 2 may best be described as representing a frequency variation whereby a first frequency (e.g., 1 Hz) may be associated with an input data bit equal to -1 and a second frequency (e.g., 2 Hz) may be associated with an input data bit equal to +1. It is to be appreciated that the GMSK output signal, ultimately transmitted by the GMSK modulator, will in effect be a frequency modulated signal whereby a transmitted signal corresponding to a -1 data bit will have the first frequency (e.g., 1 Hz) associated therewith and a transmitted signal corresponding to a +1 data bit will have the second frequency (e.g. 2 Hz) associated therewith.
However, since the GMSK modulator output signal is transmitted in the time domain, integrator 4 is required to integrate the frequency variation associated with the output signal of the Gaussian low pass filter 2 and provide the corresponding phase variation associated with a sinusoidal signal (e.g., cosine signal) varying over time at the first frequency and the corresponding phase variation associated with a sinusoidal signal (e.g., cosine signal) varying over time at the second frequency. In other words, the integrator 4 samples the output signal of the Gaussian low pass filter 2 and determines the particular phase angle for each sample period (for a given sampling rate) whereby each phase angle is associated with the sinusoidal signals respectively corresponding to a +1 data bit and a -1 data bit. For example, assuming a period and bit interval of 1 second, a 1 Hz cosine signal will linearly vary in phase from 0 degrees (at time=0) to 360 degrees (at time=T.sub.b) over such time period, while a 2 Hz cosine signal will linearly vary in phase from 0 degrees (at time=0) to 720 degrees (at time=T.sub.b) in the same time period. Therefore, assuming a sampling rate of 0.01 seconds, it is known that the phase angle associated with the 1 Hz cosine signal at t=0.01 seconds will be 3.6 degrees, while the 2 Hz cosine signal will be 7.2 degrees. Similarly, at t=0.02 seconds, the 1 Hz cosine signal will progress to a phase angle of 7.2 degrees, while the 2 Hz cosine signal will progress to a phase angle of 14.4 degrees, and so on for each successive sampling time interval.
Next, this phase angle information for each sample is presented to the signal splitter 6 in order that the phase angle information from each sample may be split into an in-phase signal and a quadrature signal. The phase angle information contained in the in-phase signal is presented to the cosine look-up table 8, while the phase angle information contained in the quadrature signal is presented to the sine look-up table 10. The cosine and sine look-up tables, 8 and 10, provide the numeric value respectively corresponding to the cosine and sine of the phase angles associated with the samples presented thereto, e.g., cos 7.2 degrees 0.9221, sin 7.2 degrees=0.1253, etc. Accordingly, a constructed cosine signal is generated from the numeric values associated with the samples and output from the cosine look-up table 8, while a constructed sine signal is likewise generated and output from the sine look-up table 10. The cosine signal represents an in-phase signal, x(t), and the sine signal represents a quadrature signal, y(t), which vary over time at the first frequency when one of the input data bits equals -1 and vary over time at the second frequency when one of the input data bits equals +1.
The in-phase signal is then mixed in the in-phase mixer 14 with the carrier frequency signal provided by the local oscillator 12, while the quadrature signal is mixed in the quadrature mixer 18 with the carrier frequency signal shifted by 90 degrees by phase shifter 16. A mixed in-phase output signal is presented to the summer 20 from the in-phase mixer 14 and a mixed quadrature output signal is presented to the summer 20 from the quadrature mixer 18. The summer 20 combines the mixed in-phase output signal and the mixed quadrature output signal to form the GMSK output signal to be transmitted.
Further, it is known that certain functions of the conventional GMSK modulator illustrated in FIG. 1 may be provided by a signal processor as disclosed in the above-mentioned article by Dr. Feher, for instance, whereby ROM (Read Only Memory) devices may be employed to provide the numeric values of the respective cosine and sine signals. However, the utilization of a signal processor with associated ROM devices in the above-described "sample by sample" approach for constructing the in-phase and quadrature modulating signals not only requires relatively large-sized ROM devices, but also disadvantageously monopolizes much of the signal processor's processing time and resources due to the fact that calculations associated with generating the cosine and sine signals must be repetitively performed for each sample. Given the fact that a relatively large number of samples must be taken per input data bit in order to accurately construct the in-phase and quadrature modulating signals, the processing efficiency of such conventional approach may be disadvantageously low.
Furthermore, the above-described conventional GMSK modulator suffers from several operational drawbacks, one such drawback being associated with the relative hardware complexity of such a GMSK modulator. While GMSK modulation, itself. provides known advantages and highly desirable features over other binary modulation approaches (e.g., increased bandwidth efficiency), the traditional GMSK modulator (see FIG. 1) requires more active components and processing stages than many other known modulators associated with alternate binary modulation techniques, e.g., QPSK. Therefore, it would be advantageous to be able to utilize the technique of GMSK modulation in a simpler, more efficient, modulator architecture as compared to those architectures previously known.